Apparatus and method for clock synchronization in a multi-point OFDM/DMT digital communications system

ABSTRACT

A multi-point communications system is set forth herein. The communications system comprises a transmitter for transmitting OFDM/DMT symbols over a predetermined number of bins across a transmission medium. The OFDM/DMT symbols are generated using at least one timing signal. At least one of the predetermined number of bins includes a pilot tone sub-symbol having a frequency corresponding to the clock signal. The communications system also includes a receiver for receiving the OFDM/DMT symbols from the transmission medium. The receiver demodulates the received symbols using at least one timing signal. The receiver has a first pilot tone search mode of operation in which the receiver adjusts its timing signal to scan the frequency range of the predetermined number of bins looking for the pilot tone sub-symbol and identifies the bin including the pilot tone sub-symbol. The receiver further has a subsequent second pilot tone acquisition mode in which the receiver adjusts the timing signal to receive the identified bin containing the pilot tone sub-symbol and measures phase differences between successive pilot tone sub-symbols to thereby perform a further adjustment of the timing signal so that the pilot tone sub-symbol is received within a frequency range sufficient for subsequent phase locked loop processing thereof.

CROSS-REFERENCE TO RELATED APPLICATIONS

[0001] The present application is a continuation-in-part application ofU.S. Ser. No. 08/700,779, filed Aug. 22, 1996, and acontinuation-in-part of U.S. Ser. No. ______, filed Apr. 24, 1997(Attorney Docket No. 11521US01).

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT

[0002] Not Applicable

BACKGROUND OF THE INVENTION

[0003] The present invention is directed to an OFDM/DMT digitalcommunications system. More particularly, the present invention isdirected to an apparatus and method for synchronizing the clocks used ina transmitter and receiver of an OFDM/DMT digital communications system.The present invention is particularly applicable in multipoint OFDM/DMTdigital communications systems.

[0004] Multi-point communications systems having a primary site that iscoupled for communication with a plurality of secondary sites are known.One such communications system type is a cable telephony system. Cabletelephony systems transmit and receive telephone call communicationsover the same cable transmission media as used to receive cabletelevision signals and other cable services.

[0005] One cable telephony system currently deployed and in commercialuse is the Cablespan 2300 system available from Tellabs, Inc. TheCablespan 2300 system uses a head end unit that includes a primarytransmitter and primary receiver disposed at a primary site. The headend unit transmits and receives telephony data to and from a pluralityof remote service units that are located at respective secondary sites.This communication scheme uses TDM QPSK modulation for the datacommunications and can accommodate approximately thirty phone callswithin the 1.9 MHz bandwidth typically allocated for suchcommunications.

[0006] As the number of cable telephony subscribers increases over time,the increased use will strain the limited bandwidth allocated to thecable telephony system. Generally stated, there are two potentialsolutions to this bandwidth allocation problem that may be usedseparately or in conjunction with one another. First, the bandwidthallocated to cable telephony communications may be increased. Second,the available bandwidth may be used more efficiently. It is oftenimpractical to increase the bandwidth allocated to the cable telephonysystem given the competition between services for the total bandwidthavailable to the cable service provider. Therefore, it is preferable touse the allocated bandwidth in a more efficient manner. One way in whichthe assigned bandwidth may be used more efficiently is to use amodulation scheme that is capable of transmitting more informationwithin a given bandwidth than the TDM QPSK modulation scheme presentlyemployed.

[0007] The present inventors have recognized that OFDM/DMT modulationschemes may provide such an increase in transmitted information for agiven bandwidth. Such systems, however, present a number of technicalproblems. One such problem is the determination of how one or moreremote receivers are to synchronize their internal clocks and timingsystems with the internal clock and timing system of a primarytransmitter at a central site. A remote receiver must first synchronizeits internal clock and timing system with the clock used by the primarytransmitter to synthesize the transmitted signal before the remotereceiver can properly demodulate the data that it receives. A furtherproblem occurs in multipoint communication systems in which there areplural groups of remote transmitters that transmit data to centrallizedtransceivers. Each group of transmitters often has its transmissionsfrequency multiplexed with transmissions from other groups before beingdemultiplexed for receipt by a particular central transceiver. Theresulting multiplexing/demultiplexing operations introduce frequencyoffsets for which compensation must be made if the receiver of thecentral transceiver is to properly extract the correct data from thesignals that is receives. The present inventors have recognized the needfor such upstream and downstream clock synchronization and havedisclosed solutions to these problems.

BRIEF SUMMARY OF THE INVENTION

[0008] A multi-point communications system is set forth herein. Thecommunications system comprises a transmitter for transmitting OFDM/DMTsymbols over a predetermined number of bins across a transmissionmedium. The OFDM/DMT symbols are generated using at least one timingsignal. At least one of the predetermined number of bins includes pilottone sub-symbols generated from a pilot tone having a frequencycorresponding to the at least one timing signal. The communicationssystem also includes a receiver for receiving the OFDM/DMT symbols fromthe transmission medium. The receiver demodulates the received symbolsusing at least one timing signal. The receiver has a first pilot tonesearch mode of operation in which the receiver adjusts its timing signalto scan the frequency range of the predetermined number of bins lookingfor the pilot tone sub-symbols and identifies the bin including thepilot tone sub-symbols. The receiver further has a subsequent secondpilot tone acquisition mode in which the receiver adjusts the timingsignal to receive the identified bin containing the pilot tonesub-symbol and measures phase differences between successive pilot tonesub-symbols to thereby perform a further adjustment of the timing signalso that the pilot tone sub-symbol is received within a frequency rangesufficient for subsequent phase locked loop processing thereof.

[0009] In accordance with one advantageous embodiment of the system, thetiming signal of the transmitter is used for timing inverse Fouriertransform processing and for carrier generation in transmitting theOFDM/DMT symbols while the timing signal of the receiver is used fortiming Fourier transform processing and for carrier generation indemodulating the received OFDM/DMT symbols.

[0010] Other features and advantages of the present invention willbecome apparent upon review of the following detailed description andaccompanying drawings.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

[0011]FIG. 1 is a schematic block diagram of a multi-pointcommunications system having a plurality of remote service unitsdisposed at a plurality of secondary sites wherein each of the remoteservice units comprises a receiver having an improved receiverarchitecture.

[0012]FIG. 2 illustrates two symbol constellations that are transmittedin two separate frequency bins in accordance with OFDM/DMT datamodulation techniques.

[0013]FIG. 3 is a block diagram of one embodiment of a head end unit anda remote service unit of the communications system of FIG. 1 showingthose components involved in downstream synchronization.

[0014]FIG. 4 illustrates various spectral distributions for the pilottone bin and adjacent bins.

[0015]FIG. 5 is a flow chart illustrating one manner of executing thefirst pilot tone search mode of receiver operation.

[0016]FIG. 6 is a flow chart illustrating one manner of executing thesecond pilot tone acquisition mode of receiver operation.

[0017]FIG. 7 is a block diagram of on embodiment of a head end unit anda remote service unit of the communications system of FIG. 1 showingthose components involved in upstream synchronization.

[0018]FIG. 8 is a flow chart illustrating one manner of executing theupstream synchronization.

DETAILED DESCRIPTION OF THE INVENTION

[0019]FIG. 1 is a block diagram of a multi-point communications systemwhich may use a remote service unit having the improved receiver andtransmitter architectures disclosed herein. As illustrated, thecommunications system, shown generally at 20 includes a head end unit(HE) 25 disposed at a primary site. The head end unit communicates witha plurality of remote service units (RSUs) 30 respectively disposed at aplurality of secondary sites, over a transmission medium 35 such as acoaxial cable.

[0020] The digital communications system 20 may, for example, be a cabletelephony system. In such an application, the head end unit 25 isdisposed at a cable television transmission facility while the remoteservice units 30 are disposed at individual customer locations, such asindividual customer homes. The transmission medium 35 would be the newor existing transmission cable used to transmit the cable televisionservices. The head end unit 25 in a cable telephony network isresponsible for communicating with and interconnecting telephone callsbetween the plurality of remote service units 30 as well ascommunicating with a central switching office 40 for sending andreceiving telephone calls from sites exterior to the local cabletelevision service area.

[0021] The present system 20 utilizes OFDM/DMT digital data modulationfor exchanging communications data between the head end unit 25 and theremote service units 30. Such OFDM/DMT digital data communicationsassign a particular amplitude, frequency, and phase for each transmitted“sub-symbol”. The transmitted “sub-symbol” represents one or moreinformation data bits that are to be transmitted between the units 25and 30. Each sub-symbol may be represented by a point within a“constellation”, the point being transmitted at a given carrierfrequency or “bin”.

[0022]FIG. 2 illustrates the use of two constellations 90 and 95, eachhaving sixteen constellation points that are capable of beingtransmitted within two separate frequency bins. As illustrated, asub-symbol having a carrier signal of frequency f, has its amplitude andphase varied depending on the constellation point that is to betransmitted. For example, a constellation point representing the binarystates 0000 is transmitted as a sub-symbol at a phase of 0, and anamplitude of A, during a designated symbol time. A constellation pointrepresenting the binary states 1111, however, is transmitted as asub-symbol at a phase of θ₂ and an amplitude of A₂ during a designatedsymbol time. Similarly, the second constellation 95, preferably havingthe same amplitude and phase designations for its sub-symbols as thefirst constellation 90, is used to modulate a second carrier frequencyf₂. The resulting modulated signals are combined into a single outputsymbol in which the individual sub-symbols are differentiated from oneanother based on their respective carrier frequencies or “bins”. It willbe recognized that many variations of the disclosed OFDM/DMTtransmission scheme are possible, the foregoing scheme being merelyillustrated herein to provide a basic understanding of OFDM/DMTcommunications.

[0023] A block diagram of one embodiment of the general components of atransmitter 97 of a head end unit 25 and a receiver 150 of a remoteservice unit 30 used for downstream clock synchronization is shown inFIG. 3. As illustrated, the transmitter 97 of the head end unit 25receives data at one or more lines 105 and supplies this data to adigital signal processor 110 and its associated components(collectively, DSP). The DSP 110 accepts the digital data and performs aFourier Transform, preferably an Inverse Fast Fourier Transform (IFFT),on the received data. The digital data resulting from the IFFTtransformation is provided to the input of a digital-to-analog converter115. The analog signal resulting from the conversion is preferablyprovided to the input of a band pass filter 120, the output of which issupplied to the input of a mixer 125 where the filtered output signal atline 130 is mixed to a first frequency f_(x). A further band pass filter135 is provided to remove the images resulting from the mixing process.The filtered output signal at line 140 is subject to a further mixing atmixer 145 where it is mixed to a second frequency f_(y) suitable fortransmission along the transmission medium 35 (e.g., RF, opticalfrequencies, etc.).

[0024] The receivers 150 of the remote service units 30 receive theOFDM/DMT data from the transmission medium 35. The received signal isdemodulated using a first frequency, preferably f_(y), at mixer 160. Themixed signal is provided to the input of a first bandpass filter 165that filters the images resulting from the mixing process. The resultingsignal is further mixed with a demodulating signal f_(x) to a basebandlevel for the receiver 150 at mixer 170. Again, the images resultingfrom the mixing process are removed by a bandpass filter 175. Thefiltered signal is applied to the input of an analog-to-digitalconverter 180 that converts the filtered analog signal to digitalsamples that are subsequently processed by the digital signal processingportions 185 of the receiver 150. In that processing, the digital datasignals received from the analog-to-digital converter 180 undergo aFourier Transform, preferably an FFT, to extract the frequency and phasecomponents of the received signal. Based on this processing, the digitalsignal processing portions 185 may reconstruct the data provided to thetransmitter 97 at lines 105 of the head end unit 25.

[0025] As illustrated in FIG. 3, the transmitter 97 of the head end unit25 includes a voltage controlled oscillator 190. The voltage controlledoscillator 190 provides a common clocking signal at lines 200 inresponse to a signal received from the DSP 110 or an analog phase lockedloop signal 192 received on a back plane. The clocking signal at lines200 is provided to, for example, the input of a digital counter 205 thatgenerates a further clocking signal at line 210 that is provided to theinput of the digital-to-analog converter 115 to control timing of theconversion process. This same clocking signal output 200 from thevoltage controlled oscillator 190 is provided to the input of frequencysynthesizer 215 and frequency synthesizer 220. The frequencysynthesizers 215 and 220 use the received clocking signal to generatethe mixing signals f_(x) and f_(y) to mixers 125 and 145, respectively.Thus, the timing of the OFDM/DMT symbol generation and the mixing of thegenerated symbols for transmission along transmission medium 35 aredependent upon the frequency and phase of the output signal 200 of thevoltage controlled oscillator 190.

[0026] As noted above in the description of the receiver 150, thereceiver 150 executes several mixing operations on the received signaland, further, performs an analog-to-digital conversion of the receivedsignal. To properly perform the mixing and conversion operations andensure the integrity of the extracted OFDM/DMT data, it is desirable tosynchronize the signals used for the generation of the OFDM/DMTtransmission by the transmitter 97 with the signals used for thedemodulation and conversion of the symbols received at the receiver 150.Applying this principal to the embodiment set forth in FIG. 3, thesignals output by frequency synthesizers 215 and 220 to mixers 125 and145 at the transmitter 97 should be synchronized with the signals outputby the frequency synthesizers 230 and 235 to mixers 160 and 170 at thereceiver 150. Similarly, the clocking signal 210 supplied to thedigital-to-analog converter 115 should be synchronized to the samplingclock signal 237 supplied to the analog-to-digital converter 180.

[0027] In the disclosed embodiment, a reference clock signal is embeddedin the baseband OFDM digital signal by the transmitter 97 in order toperform the desired synchronization. This reference clock signal takesthe form of a constant amplitude and phase sub-symbol that istransmitted in a particular frequency bin, and is called the pilot tonesub-symbol. The pilot tone sub-symbol has a frequency and phasecorresponding to the signal output 200 from the voltage controlledoscillator 190 of the transmitter 97. Preferably, several bins aredesignated to include the pilot tone sub-symbol. Each receiver 150 seeksto recover any one of these tones, with the remaining pilot tonesdesignated as backups. The receivers 150 demodulate the RF passbandsignal assigned to the RSU and attempt to extract the pilot tone fromthe baseband signal. The eventual goal is to “lock on” to the exactphase and frequency of the pilot tone. This is done by appropriatelyadjusting a voltage controlled oscillator 240 of the receiver 150 usinga phase locked loop, such as a digital phase locked loop (DPLL). Thisresults in the locking of the sampling clock signal 237 (A/D samplingrate) with the clocking signal 210 used by the transmitter 97 to controlthe timing of the D/A conversion of the binary data received at lines105. Since the RF carrier frequency and sample clocks are tied together,this circuit topology and corresponding method also simultaneouslyaccomplishes carrier recovery.

[0028] The entire pilot tone acquisition procedure can be viewed as atwo stage process comprising search and acquisition of the pilot tone.To this end, the receiver 150 operates in accordance with at least twomodes of operation. In a first pilot tone search mode of operation, thereceiver scans the frequency range of the bins transmitted by thetransmitter 97 in predetermined frequency steps looking for the bincontaining the pilot tone. Once the bin containing the pilot tonesub-symbol has been identified, the receiver 150 makes a gross timingadjustment of the output signal of voltage controlled oscillator 240 toreceive the bin including the pilot tone sub-symbol in the correctpredetermined bin location. In a subsequently occurring second pilottone acquisition mode, the receiver 150 also measures the phasedifference between consecutive pilot tone sub-symbols to adjust thetiming of the output of the voltage controlled oscillator 240 so that itis within a frequency range sufficient for subsequent phase locked loopprocessing of the pilot tone signal. After this acquisition has takenplace, the receiver 150 switches to a steady state tracking mode inwhich the phase locked loop is used to constantly maintain synchronismwith the transmitter 97.

[0029] There are at least two ways in which this subsequent synchronismmay be maintained. First, when the PLL is using the pilot-tone, theconstant pilot-tone sub-symbol is known to the PLL beforehand. For eachsymbol time, the receiver demodulates the pilot-tone bin and computesthe error between the demodulated sub-symbol and the known value. Thiserror signal then is filtered and run through the D/A to control theVCXO appropriately. In contrast, a decision directed mode may also theused to maintain synchronism. When decision directed mode is used, thePLL selects a some bin carrying random data for processing. The receiverdoes not know the transmitted sub-symbols a priori. Each symbol time,therefore, the chosen bin is demodulated and then sliced to the nearestconstellation point. The difference (or error) between the sliced andun-sliced sub-symbol is used to drive the PLL as before. Operation inthe decision directed mode is limited to situations in which thedecisions are expected to be correct.

[0030] The foregoing, two-stage operation of the receiver 150 is used at“cold start-up” of the receiver 150. This happens, for instance, when anRSU 30 powers up for the first time or when it attempts to re-establishcommunication with the HE 25 after a prolonged period of inactivity. Theacquisition process follows a successful search. However, the firstpilot tone search mode in which the receiver 150 performs the requisitesearch for the bin containing the pilot tone sub-symbol may be skippedfor “warm start-ups”, i.e. when the receiver 150 already has a fix onthe location of the pilot tone sub-symbol and is merely attempting tore-establish communication with the HE 25 after a brief period ofdisruption. After successful acquisition of the pilot tone in the secondpilot tone acquisition mode, a steady state tracking procedure isinitiated by the receiver 150 that thereafter maintains the timing ofthe RSU 30 in synchronism with the timing of the HE 25.

[0031] The foregoing modes of receiver operation are desirable for anumber of reasons. One reason relates to the result of discrepanciesbetween the output signals of the voltage controlled oscillators 190 and240 at the HE transmitter 97 and RSU receiver 150, respectively. Suchsignals potentially have frequency deviations (typically specified inppm) that, when coupled with the high RF modulating frequenciestypically used for transmission over the transmission medium 35, resultin mutual frequency offsets. For example, if the voltage controlledoscillators 190 and 240 are specified at 30 ppm and 50 ppm respectively,then for an RF modulating frequency of 750 MHz, frequency offsets up to+/−60 KHz are possible. As a result of this frequency offset, thedownconverted OFDM baseband spectrum at the receiver 150 is displaced infrequency. The frequency locking range of a typical baseband DPLL is inthe order of 500 Hz and, thus, the pilot tone will most likely lieoutside the frequency locking range of the DPLL. The foregoing receivermode operations bring the pilot tone within the locking range of such aDPLL.

[0032] As noted above, two receiver modes of operation are employed. Thefirst pilot tone search mode of operation performs a generally grossadjustment of the voltage controlled oscillator 240 that reduces thefrequency offset between the pilot tone and the output of the voltagecontrolled oscillator 240 to within a fraction of a frequency bin (e.g.,½ to ¼ of a bin). The second pilot tone acquisition mode of operationuses the gross adjustment to provide a more accurate estimate of thefrequency offset which is then used as the starting point for the DPLL.

[0033] To begin the first pilot tone search mode of operation, the RSUreceiver 150 A/D samples the baseband OFDM signal at theanalog-to-digital converter 180 and inputs the samples into, forexample, a hardware correlator 250. Details of one embodiment of such ahardware correlator, although not pertinent to the present invention,can be found in U.S. Ser. No. ______, filed Apr. 24, 1997 (AttorneyDocket No. 11521US01). The correlator 250 can be viewed as a directdigital implementation of a discrete Fourier transform (DFT) over, forexample, 9 frequency bins. In other words, the correlator 250 providessymbol rate complex outputs to the DSP 255 for subsequent processing. Inthe present embodiment, a substantial portion of the first and secondreceiver modes of operation are implemented in the DSP 255 on symbolrate outputs from the hardware correlator 250.

[0034] In accordance with one embodiment of the first pilot tone searchmode of receiver operation, the number of bins that need to be searchedhas to be determined. Generally, such a determination depends upon theparticular system requirements. For exemplary purposes, it will beassumed that the system employs a bin width of 9.615 KHz and uses apilot tone bin that is located as the 12^(th) bin in a total receivewindow of 26 bins. The number of bins to be searched depends on theaccuracy of the voltage controlled oscillators 190 and 240 at the HE 25and RSU 30. If frequency offsets of up to +/−60 KHz are possible (seeabove), and each bin is 9.615 KHz wide, 13 bins will be searched—the 12′bin and 6 bins on each side of the 12^(th) bin in the window.

[0035] It is also desirable to select a predetermined spectral patternthat will be used to find the pilot tone. As such, a sufficient andminimal set of metrics based on the pattern that will indicate with highprobability that the pilot is found may be employed in the first pilottone search mode. This pattern should be limited to as few bins aspossible to minimize bandwidth usage while at the same time beingdistinct from the rest of the spectrum. Several possible symmetric andasymmetric spectral patterns are shown in FIGS. 4(a)-4(d).

[0036] For exemplary purposes, the 5-bin-wide pattern of FIG. 4(a) isemployed in the instant case. This pattern has a pilot tone disposed inone bin with two adjacent empty (zero power) bins on each side thereof.The two outermost bins which are beside each of the empty bins carryeither actual data or random data. This ensures that there will alwaysbe a stable non-zero average power in these two outermost bins. Notethat the wider the pattern is, the larger the number of bins that needto be received. For instance, if 13 bins are possible candidates for thepilot location, then this 5-bin pattern requires that the receiver 150receive and process 17 bins.

[0037] It is further assumed for purposes of the present example thatthere are eight, 26-bin windows transmitted by the HE 25, and that fourpredetermined windows among the eight will include predetermined binshaving the spectral pattern with the pilot tone. Pursuant to the firstpilot tone search mode of operation, the receiver 150 of the RSU 30adjusts to receive one of the predetermined windows with the pilot tone.A selected window number is predetermined and back-up windows are alsopredetermined in the event that pilot recovery fails for the window ofchoice.

[0038]FIG. 5 illustrates one manner of implementing the first pilot tonesearch mode in the receiver 150. In the specific implementationillustrated here, a wide range of tests are performed on receivedsignals to verify the presence of the pilot tone. However, it will berecognized that fewer than all such tests may be used in this mode yetstill achieve satisfactory results.

[0039] In connection with the search mode, each RSU 30 has a table of,for example, five frequency offset values (also called DAC offset table)for its voltage controlled oscillator 240 for each window. The DAC tableis used to scan a window in, for example, five steps of 1.923 KHz each(9615/5 Hz). By using scanning steps equal to a fifth of the width of abin, it is possible get an estimate of the pilot location to within1.923 KHz (9615/5 Hz). To implement the scanning, the receiver 150 ofthe RSU 30 runs through a table of five output values supplied to thedigital-to-analog converter 265. The output of the digital-to-analogconverter 265 is supplied to the input of a low pass filter 270, theoutput of which is an analog control voltage that alters the frequencyof the output signal from the voltage controlled oscillator 240. Thus,each value that is output corresponds to a frequency offset of thevoltage controlled oscillator 240. For each offset value, the RSU 30carries out a number of functions.

[0040] First, the receiver 150 changes the frequency offset of thevoltage controlled oscillator 240 by a fifth of the width of a bin. Thischange of the frequency offset is performed by writing the correct DACoutput value in the table of five DAC output values to thedigital-to-analog converter 265. Immediately after writing to thedigital-to-analog converter 265, several symbols have to be discarded toallow the frequency output of the voltage controlled oscillator 240 tostabilize. The number of symbols to be discarded will depend on thevoltage controlled oscillator properties and the magnitude of the changein frequency. As such, the receiver 150 may need to discard a fewhundred symbols.

[0041] Next, a complex phase correction term due to the frequency offsetin the receiver 150 is applied to each received sub-symbol. Such a phasecorrection is necessary when the data transmitted by the transmitter 97is in the form of a formatted data frame having, for example, a cyclicprefix or an analogous counterpart. The phase is corrected in accordancewith the following equation:

J _(i, j)(l)=I _(i, j)(l)e ^({square root}{square root over (−1)}Φl)

[0042] For i=1, . . . , 17 and j=0,1,2,3,4 and where$\Phi = {2\pi \quad {f_{off}\left( \frac{CP}{N} \right)}}$

[0043] is the incremental phase correction applied to all receivedsub-symbols in the chosen window received with an offset of f_(off)frequency bins with respect to the transmitted bins (see the discussionin the following paragraph), and 1 is the symbol index. Here, the jindex corresponds to the index into the DAC offset table correspondingto the frequency steps used to scan the predetermined window having thepilot tone. The term I_(i,j)(l) is the received symbol in the i^(th) binfor the j^(th) DAC offset table value.

[0044] This incremental phase shift is applied in systems in which acyclic prefix is used and each of the remote receivers are designed toonly process signals within a passband beginning at a fixed frequencycorresponded to a predetermined bin (e.g., bin 64). In such a system,the HE transmits across substantially the entire available signalspectrum. However, the receiver of each RSU mixes the transmitted signalso that the frequency range of the window that it is to process is mixedto begin at the fixed frequency corresponding to the predetermined bin.The f_(off) values for each window, accordingly, are system dependentand, thus, are dependent on the system design parameters. It should benoted that the value of CP is zero if a cyclic prefix, or an analogouscounterpart, is not transmitted from the transmitter 97.

[0045] After the phase correction has been completed, the power of thesignal received in each of the bins is computed as is the signal'scorrelation. Assuming that the pilot signal bin is known to be the 12thbin in the window, the squared magnitude of each symbol in the 12th binin the window and the 8 bins on either side of the 12th bin (for a totalof 17 bins) is computed.

[0046] The receiver 150 computes the correlation of the current symbolJ_(i,j)(l)with the complex conjugate of the previous symbol J_(i,j)(l−1)for the 12th bin and 6 bins on each side of the 12th bin (for a total of13 bins) in accordance with the following equation:

R _(i, j)(l)=J _(i, j)(l)×J _(i, j)*(l−1)

[0047] where i is the bin number and j is the index into the DAC offsettable for the voltage controlled oscillator 240 for each of theseequations. The correlation will be constant if the bin number and indexincludes the pilot tone.

[0048] The foregoing power and correlation calculations are thenaveraged for each bin i and index j over L symbols. Accordingly, theforegoing operations are repeated for L symbols to compute the averagesin accordance with the following averaging equations:${P_{i,j}(L)} = {\frac{1}{L}{\sum\limits_{l = 1}^{L}{{J_{i,j}(l)}}^{2}}}$${H_{i,j}(L)} = {{\frac{1}{L}{\sum\limits_{l = 1}^{L}{R_{i,j}(l)}}}}^{2}$

[0049] where P_(i,j) and H_(i,j) are the average power and coherence inthe ith bin and for the jth index over L symbols.

[0050] After averaging, the metrics for each bin i and index j arecomputed. To this end, for each index j, the following metrics arecomputed for the 12th bin and 6 bins on each side of the 12th bin (for atotal of 13 bins):

A _(i,j) =P _(i,j)(L)−P _(i−1,j)(L)−P _(i+1,j)(L)

B_(i,j) =|P _(i−1,j)(L)−P _(i+1,j)(L)|

C_(1, i,j) =P _(i−2,j)(L)−P _(i−1,j)(L)

C_(2 i,j) =P _(i+2,j)(L)−P _(i+1,j)(L)

[0051] Note that the number of metrics and the definition of the metricswill depend on the spectral patterns used, the foregoing metrics beingthose used for the spectral pattern of FIG. 4(a).

[0052] If the table of output values to the voltage controlledoscillator 240 is not exhausted after computing the metrics, the entireprocess above is repeated using the next value in the DAC offset table.If the table has been exhausted, H_(i,j)(L) and P_(i,j)(L) arere-ordered in a predetermined fashion. Where the DAC offset index runsfrom 0 to 4, H_(i,j)(L) is re-ordered as H_(i,0)(L), H_(i,l)(L),H_(i,2)(L), . . . The same re-ordering is performed on array P_(i,j)(L).Such ordering can be incorporated into earlier operation to saveprocessing time by using appropriate addressing methods. After thisrearrangement, each of the arrays should have 65 elements.

[0053] The square root of each element H_(i,j)(L) in the array ofcorrelation metrics is then computed and the result is divided by thecorresponding average power value P_(i,j)(L) to provide a normalizedcorrelation metric T_(i,j) in accordance with the following equation:$T_{i,j} = \frac{\sqrt{H_{i,j}(L)}}{P_{i,j}(L)}$

[0054] The pilot tone will have a high degree of coherence and hence theT_(i,j) values will be large (close to 1) for bins in the neighborhoodof the bin containing the pilot tone. All other bins will have very lowcoherence (close to 0) as they will either be carrying uncorrelated dataor will have zero power and be effectively Gaussian noise. Thus, thecoherence test is a simple threshold test which eliminates fromconsideration all bins that have T_(i,j) values less than apredetermined value. This predetermined value, for example, may be inthe range of 0.5-0.8. The test can thus be represented as follows:

[0055] Reject (bin, index) pair (i,j) if T_(i,j)<λ where 0.5<λ<0.8.

[0056] If all the bins in the window are eliminated, then the receiver150 will move on to a backup window and repeat the foregoing searchprocess.

[0057] In addition to undergoing a coherence test, the acquired signalsmay also undergo an excessive coherence test that is executed during thefirst pilot tone search mode. This test is used to reject a window ifthere are coherent interferers too close to the pilot tone and/or toomany coherent interferers in the window. This test may be used tosuccessfully reject video channels when hands-off provisioning isdesired. In NTSB standard video channels, the spectrum has severalcarriers that are spaced evenly at about 16 KHz apart. This is veryclose to the bin spacing that is used in the exemplary embodiment here.If only the foregoing coherence test and the pattern matching tests(described below) were used, several bin groupings would pass thesetests. Thus, the test for excessive coherence is useful to ensure thatthere is only one grouping of bins that passes the tests.

[0058] The excessive coherence test involves using a lowpass filterwith, for example, 13 taps with all taps set to unity to filter thearray of T_(i,j) values. The maximum output of the filter, T_(max), iscompared to a threshold value T_(th), that is determined, for example,experimentally. If T_(max)<T_(th), then the window is rejected and thereceiver will move on to the next window and begin processing anew. Notethat the evenly spaced spectral pattern of the cable channel could berejected with either of the asymmetric spectral patterns in FIGS.4(c)-4(d) with appropriate pattern matching tests. However, theexcessive coherence test is still useful in the case of coherentinterference near the pilot tone.

[0059] The receiver 150 may then perform one or more successive patternmatching tests. The pattern matching tests compare the spectral patternaround each bin with the expected pattern in FIG. 4(a). The particularpattern matching tests depend on which pattern is used. In accordancewith a first test, the receiver rejects(bin,index) pairs that satisfythe following criterion:

C_(1, i,j)<0 or C_(2, i,j)<0

[0060] Another possible rejection criterion that can be used instead ofthe above is

C_(1, i,j)<B_(i,j) or C_(2, i,j)<B_(i,j)

[0061] Again, if all the bins in the window are rejected, then thereceiver will move on to a backup window and start anew.

[0062] A further subsequent pattern matching test may also be performedby the receiver 150. In accordance with this further test, the receiverrejects the bin/index pairs that satisfy the following criterion:

A_(i, j)<μB_(i,j)

[0063] The value of μ is, for example, selected experimentally andusually lies between 1 and 10. If all the bins in the window areeliminated, then the receiver will move on to a backup window and startanew.

[0064] After the above tests are performed, all the bins except for afew grouped around the bin containing the pilot tone will be rejected.The best bin i and index j into the DAC offset table for the voltagecontrolled oscillator 240 is found by searching for the largest T_(i,j)value among the 65 values. Using the index j corresponding to thelargest T_(i,j), the output value that is to be provided to thedigital-to-analog converter 265 is found from the table of DAC outputvalues and the voltage-controlled-oscillator is directed to proceed tothe correct position to allow further processing in the second pilottone acquisition mode of operation.

[0065] One embodiment of the implementation of the second pilot toneacquisition mode is set forth in connection with FIG. 6. With respect tothe second pilot tone acquisition mode, it can be shown that in theabsence of any interference from data carrying bins, there is a phaseoffset (in radians) between$\Theta = {{2{\pi \left( {ɛ + {\left( {f_{off} + ɛ} \right)\left( \frac{CP}{N} \right)}} \right)}} = {{2{{\pi ɛ}\left( {1 + \frac{CP}{N}} \right)}} + \Phi}}$

[0066] consecutive pilot symbols given by the equation: where ε is thefractional (normalized) frequency offset in bins. For example, if theresidual frequency offset after the first pilot tone search mode iscompleted is 1 KHz, and for a bin width of 9.6 KHz, ε=1/9.6=0.104. Thephase rotation Φ is predetermined by the window number and is correctedas soon as the correlator outputs are available to the DSP. In thesecond pilot tone acquisition mode, the receiver 150 extracts the phasedifferences between consecutive pilot tone sub-symbols to estimate ε. Tothis end, let I(n) denote the nth complex symbol obtained after thephase correction noted above. To estimate the frequency offset, thereceiver 150 forms a sum using N symbols as,${S(N)} = {\sum\limits_{l = 1}^{N}{{J(l)}{J^{*}\left( {l - 1} \right)}}}$

[0067] where J*(n) denotes the complex conjugate of the nth symbol. Anestimate of the fractional frequency offset is then obtained as$\hat{ɛ} = {\frac{1}{2{\pi \left( {1 + \frac{CP}{N}} \right)}}{{atan}\left( \frac{{Im}\left( {S(N)} \right)}{{Re}\left( {S(N)} \right)} \right)}}$

[0068] Since {circumflex over (ε)} is a signed dimensionless numbernormalized by the bin width, it is converted to a frequency offset (inHertz) by multiplying it by the bin width. This frequency offset istranslated to the appropriate numerical value and written out to thedigital-to-analog converter 265. To meaningfully interpret the output ofthe second pilot tone acquisition mode, |{circumflex over (ε)}|<0.5,i.e. the magnitude of the residual frequency offset is less than half ofa bin width.

[0069] After the receiver 150 has completed operation in the secondpilot tone acquisition mode, the voltage controlled oscillator 240 isset to the best estimate of the pilot tone and is within the lockingbandwidth of a DPLL. The DPLL begins operation by utilizing thedifference between the phase of the received pilot tone and its constantdesired (transmitted) value as the error signal. This symbol rate errorsignal is filtered by, for example, a type II proportional-integral (PI)loop filter to generate the control voltage for adjusting the voltagecontrolled oscillator 240. The second order loop filter ensures a zerosteady state error in the face of frequency offsets. The loop filtertransfer function is given by,${L(z)} = {K_{P} + \frac{{zK}_{l}}{z - 1}}$

[0070] which results in an overall transfer function of the form.${H(z)} = \frac{{\left( {K_{P} + K_{l}} \right)z} - K_{P}}{z^{2} + {\left( {K_{P} + K_{l} - 2} \right)z} + 1 - K_{P}}$

[0071] With appropriately chosen DPLL parameters, phase lock can beachieved within a few thousand symbols. Also, several sets of DPLLparameters are chosen to provide multiple loop bandwidths. For instance,the loop bandwidth takes on the highest value during acquisition and thelowest during tracking.

[0072] After sample timing has been acquired, the RSU receiver 150constantly maintains the correct timing. This requires the continuousoperation of the DPLL. Since each RSU 30 in the present embodiment candemodulate at most 9 bins, it is wasteful of bandwidth to constantly usethese bins for a pilot tone. Thus, after successful acquisition, theDPLL preferably switches to any one of the data carrying bins andoperates in a decision directed tracking mode. For relatively cleandownstream channels, this approach is robust and bandwidth conserving.

[0073] From the foregoing description of the clock synchronization, itis seen that all RSUs synchronize their receive clocks to the masterclock provided by the HE. The transmit clock used to synthesize thetransmissions from the transmitter of each RSU 30 is preferably likewisegenerated from the voltage controlled oscillator 240 at the RSU 30.Similarly, the receive clocks of the receiver 287 (see FIG. 7) of the HE25 may be generated from voltage controlled oscillator 190. The XIMTVCXO 190 itself is derived, for example, via an analog-PLL signal that,for example, is received from a HE backplane. Thus, the HE 25 need onlyprovide a stable clock reference and the burden of synchronizationlargely rests with the RSUs 30.

[0074] This situation, however, is altered by any frequency shiftsincurred in the upstream transmissions from the RSUs 30 to the HE 25.Specifically, in some multipoint communication systems, frequencyoffsets are introduced due to multiplexing of upstream signals fromseveral remote sites onto a single optical fiber at intermediate nodes,for example, electro-optical nodes. The signals are demultiplexed at theHE 25 and downconverted to the original sub-split return frequency band.Use of such conversions can greatly reduce the noise level on theupstream channel along the transmission medium 35. However, the clocksused for multiplexing and de-multiplexing at these nodes are typicallynot synchronized with each other (i.e. they are relatively free running)and introduce frequency offsets, possibly in the range of several KHz.Such a system is illustrated in FIG. 7.

[0075] The present inventors have recognized this problem and haveprovided a solution. With reference to FIG. 7, a second voltagecontrolled oscillator 290 is used at the HE 25 which serves as the clockreference for one of the frequency synthesizers 295 included in the RFsub-system of the receiver 287 of the HE 25. The voltage controlledoscillator 290 has its output signal at line 300 synchronized with apilot tone provided by the transmitter 277 of the RSU 30. The pilot tonefrom the RSU 30 is provided in a predetermined bin. The HE 25 and RSU 30need only carry out the synchronization process once when the first RSUpowers up and is seeking to establish upstream communication with the HE25 or after bi-directional communications have been disrupted for aprolonged interval of time. Thereafter, the HE 25 maintains synchronismwith all RSU transmitters 277 by operating a symbol rate acquisitionDPLL in decision directed tracking mode. The use of this separatelysynchronized voltage controlled oscillator 290 compensates for frequencyoffsets introduced by the free running clocks of the multiplexers 310and demultiplexer 315. Note that in-spite of the upstream frequencyoffsets, the analog-to-digital converter 320 of the receiver 287 of theHE 25 and frequency synthesizer 325 used by mixer 30 can still be drivenoff the voltage controlled oscillator 190.

[0076] The upstream synchronization process is initiated after the RSU30 has acquired downstream synchronization and thereby has thecapability to properly receive messages from the HE 25. Concurrent withthe transmission of pilot tones, the HE demodulates all bins andcontinually measures the received power in a predetermined window of apredetermined number of bins (possibly all bins designated forcommunication) around a pre-designated bin. This bin, designated as theupstream pilot tone bin, is used by the first RSU 30 to transmit anupstream pilot tone after deriving its own upstream clock via thesynchronization process described above. Since there is only onetransmitting RSU 30, the HE 25 can locate the pilot tone bin withrelative ease, compute the frequency offset (if any), and appropriatelyadjust voltage controlled oscillator 290.

[0077] One embodiment of the upstream synchronization process is setforth in connection with FIG. 8. As illustrated, the synchronizationprocess begins with a search for the pilot tone transmitted by the RSU30. Pursuant to the synchronization process, an initialization is firstperformed. In the initialization process, the transmitter 277 of the RSU30 transmits a pilot tone having constant phase and constant amplitudeafter it has successfully executed the downstream pilot tone search andacquisition processes. The pilot tone is transmitted in the upstreampilot tone bin for a predetermined number of consecutive symbols unlessit receives a message from the HE 25 to do otherwise.

[0078] If upstream synchronization is to proceed after initialization,the HE 25 demodulates all bins and continuously measures receive powerin a window of a predetermined number of bins centered around thepre-designated upstream pilot tone bin. The HE 25 then locates the binfor which (a) receive power is maximum and (b) the magnitude ofdifference in power with adjacent bins exceeds a predeterminedthreshold. Such determinations can be made in the manner describedabove. Additionally, further coherency tests, etc., may also optionallybe employed. This bin is selected as the bin which is most likely tocontain the pilot tone. The HE 25 then uses the selected bin to computethe frequency offset (in bins) of the selected peak power bin from itsexpected location. Note that the offset would be zero when themuxing/demuxing process is ideal. The voltage controlled oscillator 290is then adjusted to compensate for the frequency offset by translatingthe offset frequency into a signed control voltage that is applied tothe voltage controlled oscillator 290 by a digital-to-analog converter330 (and, preferably, through a low pass filter 335) that is responsiveto a control signal provided by the digital signal processor 340. The HE25 then waits for a predetermined number of symbols and further refinesany adjustment that may be necessary until a zero offset results.

[0079] After the upstream pilot tone search process has beensuccessfully completed, the HE 25 executes an upstream acquisitionprocess. This acquisition process is substantially similar to theprocess noted above in connection with the second pilot tone acquisitionmode of the RSU receiver 150. The DPLL parameters for the HE 25,however, are likely to be different than those used in the RSU DPLL.After acquisition is complete, the HE 25 may send an appropriate messageto the first RSU to indicate successful capture of and synchronizationwith the upstream pilot tone. At this point, the HE 25 and RSU 30 may beconsidered to have successfully carried out sample timing and carrierrecovery for both the downstream and upstream channels.

[0080] After successful acquisition has taken place, the HE 25 needs toconstantly run the acquisition DPLL in a tracking mode. Loss of trackingcan lead to disruption in upstream communication with the entire RSUpopulation (often several hundred RSUs) served by the HE 25. Unlikeacquisition, tracking can reliably proceed in decision directed mode.The HE 25 instructs the first RSU to transmit random data in a bin lyingwithin the transmit range of the RSU 30. Even though the HE 25 employsrandom data transmissions from any one RSU 30, it can designate morethan one RSUs for additional reliability. This process continuesunchanged even if the RSUs being used for tracking begin to carry livetraffic.

[0081] Other aspects of an OFDM/DMT communications system are set forthin U.S. Ser. No. ______, filed Apr. 24, 1997, titled “SYMBOL ALIGNMENTIN A MULTIPOINT OFDM/DMT DIGITAL COMMUNICATION SYSTEM” (Attorney DocketNo. 11521US01) and in co-pending application (Attorney Docket No.11824US01), titled “APPARATUS AND METHOD OR UPSTREAM CLOCKSYNCHRONIZATION IN A MULTIPOINT OFDM/DMT DIGITAL COMMUNICATION SYSTEM”,that are hereby incorporated by reference. These other aspects, however,are not particularly pertinent to the present synchronization system.

[0082] Although the present invention has been described with referenceto specific embodiments, those of skill in the art will recognize thatchanges may be made thereto without departing from the scope and spiritof the invention as set forth in the appended claims.

1. A communications system comprising: a transmitter for transmittingOFDM/DMT symbols over a predetermined number of bins across atransmission medium, the OFDM/DMT symbols being generated using at leastone timing signal, at least one of the predetermined number of binsincluding a pilot tone sub-symbol having a frequency corresponding tothe sampling clock signal; a receiver for receiving the OFDM/DMT symbolsfrom the transmission medium, the receiver demodulating the receivedsymbols using at least one timing signal, the receiver having a firstpilot tone search mode of operation in which the receiver adjusts itstiming signal to scan the frequency range corresponding to thepredetermined number of bins looking for the pilot tone sub-symbol andidentifies the bin including the pilot tone sub-symbol, the receiverfurther having a subsequent second pilot tone acquisition mode in whichthe receiver adjusts its timing signal to receive the identified bincontaining the pilot tone sub-symbol and measures phase differencesbetween successive pilot tone sub-symbols to thereby perform a furtheradjustment of the timing signal so that the pilot tone sub-symbols arereceived within a frequency range sufficient for subsequent phase lockedloop processing thereof.
 2. A communications system as claimed in claim1 wherein the timing signal of the transmitter is used for timinginverse Fourier transform processing and for carrier generation intransmitting the OFDM/DMT symbols.
 3. A communications system as claimedin claim 1 wherein the timing signal of the receiver is used for timingFourier transform processing and for carrier generation in demodulatingthe received OFDM/DMT symbols.
 4. A communications system as claimed inclaim 2 wherein the timing signal of the receiver is used for timingFourier transform processing and for carrier generation in demodulatingthe received OFDM/DMT symbols.
 5. A communications system as claimed inclaim 1 wherein the receiver computes coherency of sub-symbols receivedin each bin scanned during the first pilot tone search mode ofoperation, the receiver rejecting bins having sub-symbols with coherencybelow a predetermined threshold as not including the pilot tonesub-symbols.
 6. A communications system as claimed in claim 1 whereinthe receiver computes excessive coherency of sub-symbols received ineach bin having a coherency above the predetermined threshold, thereceiver rejecting bins having sub-symbols having excessive coherency asnot including the pilot tone sub-symbols.
 7. A communications system asclaimed in claim 1 wherein the bin including the pilot tone sub-symbolis disposed proximate a plurality of proximate bins, the bin containingthe pilot tone sub-symbol and the plurality of proximate bins defining apredetermined spectral pattern.
 8. A communications system as claimed inclaim 1 wherein the receiver identifies the known spectral pattern whilein the first pilot tone search mode of operation to assist inidentifying the frequency of the bin containing the pilot tonesub-symbol.
 9. A communications system as claimed in claim 8 wherein thespectral pattern is symmetric.
 10. A communications system as claimed inclaim 8 wherein the spectral pattern is asymmetric.
 11. A communicationssystem as claimed in claim 1 wherein the OFDM/DMT symbols aretransmitted in periodically occurring formatted symbol frames, eachformatted symbol frame having a cyclic prefix.
 12. A communicationssystem as claimed in claim 11 wherein the receiver adjusts the timingsignal in the second mode of operation in accordance with a plurality ofoperations, the plurality of operations comprising: computing a sum S(N)of N consecutive pilot tone sub-symbols in accordance with the followingequation${S(N)} = {\sum\limits_{l = 1}^{N}{{J(l)}{J^{*}\left( {l - 1} \right)}}}$

where J(n+1) denotes the (n+1)th complex pilot tone sub-symbol obtainedafter phase correction for the cyclic prefix and J*(n) is the complexconjugate of the previously obtained complex pilot tone sub-symbol afterphase correction for the cyclic prefix; obtaining an estimate of anormalized frequency offset ε in accordance with the following equation$\hat{ɛ} = {\frac{1}{2{\pi \left( {1 + \frac{CP}{N}} \right)}}{{atan}\left( \frac{{Im}\left( {S(N)} \right)}{{Re}\left( {S(N)} \right)} \right)}}$

where CP is the duration of the cyclic prefix in samples; using thenormalized frequency offset to calculate the total frequency offset usedto adjust the output of the voltage controlled oscillator.
 13. Acommunications system as claimed in claim 1 wherein the receiver adjuststhe timing signal in the second mode of operation in accordance with aplurality of operations, the plurality of operations comprising:computing a sum S(N) of N consecutive pilot tone sub-symbols inaccordance with the following equation${S(N)} = {\sum\limits_{l = 1}^{N}{{J(l)}J*\left( {l - 1} \right)}}$

where J(n+1) denotes the (n+1)th complex pilot tone sub-symbol obtainedafter phase correction for the cyclic prefix and J*(n) is the complexconjugate of the previously obtained complex pilot tone sub-symbol afterphase correction for the cyclic prefix; obtaining an estimate of anormalized frequency offset E in accordance with the following equation$\hat{ɛ} = {\frac{1}{2\pi}{{atan}\left( \frac{{Im}\left( {S(N)} \right)}{{Re}\left( {S(N)} \right)} \right)}}$

using the normalized frequency offset to calculate the total frequencyoffset used to adjust the output of the voltage controlled oscillator.14. A communications system comprising: a transmitter for transmittingOFDM/DMT symbols over a predetermined number of bins across atransmission medium, the OFDM/DMT symbols being generated using a timingclock signal, at least one of the predetermined number of bins includinga pilot tone sub-symbol having a frequency corresponding to the timingclock signal; a receiver for receiving the OFDM/DMT symbols from thetransmission medium, the receiver comprising a demodulator for receivingthe OFDM signals at a selectable reception frequency determined by afrequency control signal and for converting received OFDM/DMT symbols todigital data samples in accordance with a sampling clock signal, thesampling clock signal having a frequency that is responsive to asampling clock control signal, a controller circuit including a firstoutput signal connected to provide the frequency control signal to thedemodulator circuit and a second output signal connected to provide theclock control signal to the demodulator, the controller including afirst pilot tone acquisition mode of operation in which the clockcontroller (i) directs the demodulator to scan at least a portion of theplurality of bins by varying the frequency control signal to thedemodulator, (ii) uses the digital data samples from the demodulator toidentify which of the plurality of bins includes the pilot tone, thecontroller further including a second pilot tone acquisition mode inwhich the controller (i) provides a frequency control signal to thedemodulator to direct the demodulator to receive the bin including thepilot tone as identified in the first pilot tone search mode, and (ii)uses the digital data samples from the demodulator to provide afrequency control signal to the demodulator so that the demodulatorreceives the bin containing the pilot tone within a frequency rangesufficient for subsequent phase locked loop processing of the pilot tonewhereby the frequency and phase of the sampling clock signal of thereceiver is responsive to the phase locked loop processing to therebysynchronize the sampling clock of the receiver with the timing clocksignal of the transmitter.
 15. A communications system as claimed inclaim 14 wherein the demodulator comprises: a voltage controlledoscillator responsive to the frequency control signal, the voltagecontrolled oscillator having an output signal of variable frequency thatis adjustable in response to the frequency control signal; and afrequency synthesizer responsive to the output signal of the voltagecontrolled oscillator for generating a demodulating carrier signal thatdetermines the selectable reception frequency.
 16. A communicationssystem as claimed in claim 14 wherein the demodulator comprises: avoltage controlled oscillator responsive to the sampling clock controlsignal, the voltage controlled oscillator having an output signal ofvariable frequency that is adjustable in response to the sampling clockcontrol signal; and a sampling clock signal generator for generating thesampling clock in response to the output signal of the voltagecontrolled oscillator.
 17. A communications system as claimed in claim14 wherein the demodulator comprises: a voltage controlled oscillatorresponsive to the frequency control signal, the voltage controlledoscillator having an output signal of variable frequency in response tothe frequency control signal; a frequency synthesizer responsive to theoutput signal of the voltage controlled oscillator for generating ademodulating signal that determines the selectable reception frequency;a clock signal generator for generating the sampling clock in responseto the output signal of the voltage controlled oscillator; and thesampling clock control signal and the frequency control signal being acommon signal.
 18. A communications system as claimed in claim 14wherein the controller uses the digital data samples of the demodulatorto compute squared magnitude values to identify the bin containing thepilot tone while in the first pilot tone acquisition mode of operation.19. A communications system as claimed in claim 14 wherein thecontroller uses the digital data samples of the demodulator to computecoherency of symbols received in each bin of the at least one portion ofthe plurality of bins that are scanned during the first pilot toneacquisition mode of operation, the controller rejecting bins havingcoherency below a predetermined threshold as not including the pilottone.
 20. A communications system as claimed in claim 14 wherein thecontroller uses the digital data samples of the demodulator to computeexcessive coherency of symbols received in each bin having a coherencyabove the predetermined threshold, the controller rejecting bins havingexcessive coherency as not including the pilot tone.
 21. Acommunications system as claimed in claim 14 wherein the bin includingthe pilot tone is disposed proximate a plurality of proximate bins, thebin containing the pilot tone and the plurality of proximate binsdefining a predetermined spectral pattern.
 22. A communications systemas claimed in claim 14 wherein the controller uses the digital datasamples of the demodulator to identify the known spectral pattern whilein the first pilot tone acquisition mode of operation.
 23. Acommunications system as claimed in claim 22 wherein the spectralpattern is symmetric.
 24. A communications system as claimed in claim 22wherein the spectral pattern is asymmetric.
 25. A communications systemas claimed in claim 14 wherein the controller uses the digital data ofthe demodulator to calculate phase differences between consecutive pilottone sub-symbols to adjust the demodulator in the second mode ofoperation.
 26. A communications system as claimed in claim 14 whereinthe OFDM/DMT symbols are transmitted in periodically occurring formattedsymbol frames, each formatted symbol frame having a cyclic prefix.
 27. Acommunications system as claimed in claim 26 wherein the controller usesdigital data samples of the received OFDM/DMT signals to adjust thedemodulator in the second mode of operation in accordance with aplurality of operations on the digital data samples, the plurality ofoperations comprising: computing a sum S(N) of N consecutive pilot tonesub-symbols in accordance with the following equation${S(N)} = {\sum\limits_{l = 1}^{N}{{J(l)}J*\left( {l - 1} \right)}}$

where J(n+1) denotes the (n+1)th complex pilot symbol obtained afterphase correction for the cyclic prefix and J*(n) is the complexconjugate of the previously obtained complex pilot symbol after phasecorrection for the cyclic prefix; obtaining an estimate of a normalizedfrequency offset E in accordance with the following equation$\hat{ɛ} = {\frac{1}{2{\pi \left( {1 + \frac{CP}{N}} \right)}}{{atan}\left( \frac{{Im}\left( {S(N)} \right)}{{Re}\left( {S(N)} \right)} \right)}}$

where CP is the duration of the cyclic prefix in bins; using thenormalized frequency offset to calculate the total frequency offset usedto adjust the output of the voltage controlled oscillator.
 28. Acommunications system as claimed in claim 14 wherein the controller usesdigital data samples of the received OFDM/DMT signals to adjust thedemodulator in the second mode of operation in accordance with aplurality of operations on the digital data samples, the plurality ofoperations comprising: computing a sum S(N) of N consecutive pilot tonesub-symbols in accordance with the following equation${S(N)} = {\sum\limits_{l = 1}^{N}{{J(l)}J*\left( {l - 1} \right)}}$

where J(n+1) denotes the (n+1)th complex pilot symbol obtained afterphase correction for the cyclic prefix and J*(n) is the complexconjugate of the previously obtained complex pilot symbol after phasecorrection for the cyclic prefix; obtaining an estimate of a normalizedfrequency offset {circumflex over (ε)} in accordance with the followingequation$\hat{ɛ} = {\frac{1}{2\pi}{{atan}\left( \frac{{Im}\left( {S(N)} \right)}{{Re}\left( {S(N)} \right)} \right)}}$

using the normalized frequency offset to calculate the total frequencyoffset used to adjust the output of the voltage controlled oscillator.29. A communications system as claimed in claim 14 wherein thecontroller comprises a digital signal processor.
 30. A communicationssystem comprising: a transmitter for transmitting OFDM/DMT symbols overa predetermined number of bins across a transmission medium, theOFDM/DMT symbols being generated using at least one timing signal, atleast one of the predetermined number of bins including a pilot tonehaving a frequency corresponding to the sampling clock signal; areceiver for receiving the OFDM/DMT symbols from the transmissionmedium, the receiver demodulating the received symbols using at leastone timing signal, the receiver having a first pilot tone search mode ofoperation in which the receiver adjusts its timing signal to scan thepredetermined number of bins looking for the pilot tone and identifiesthe bin including the pilot tone, the receiver further having asubsequent second pilot tone acquisition mode in which the receiveradjusts its timing signal to receive the identified bin containing thepilot tone and performs a further adjustment of the timing signal sothat the pilot tone is received within a frequency range sufficient forsubsequent phase locked loop processing thereof.